Quadrature Amplitude Modulation
Because of the orthogonality property of the carrier signals, it is possible to detect the modulating signals independently. In the ideal case I(t) is demodulated by multiplying the transmitted signal with a cosine signal: [pic] Using standard trigonometric identities, we can write it as: [pic] Low-pass filtering ri(t) removes the high frequency terms (containing 4? f0t), leaving only the I(t) term. This filtered signal is unaffected by Q(t), showing that the in-phase component can be received independently of the quadrature component.
Similarly, we may multiply s(t) by a sine wave and then low-pass filter to extract Q(t). The phase of the received signal is assumed to be known accurately at the receiver. If the demodulating phase is even a little off, it results in crosstalk between the modulated signals. This issue ofcarrier synchronization at the receiver must be handled somehow in QAM systems. The coherent demodulator needs to be exactly in phase with the received signal, or otherwise the modulated signals cannot be independently received.
For example analog television systems transmit a burst of the transmitting colour subcarrier after each horizontal synchronization pulse for reference. Analog QAM is used in NTSC and PAL television systems, where the I- and Q-signals carry the components of chroma (colour) information. “Compatible QAM” or C-QUAM is used in AM stereo radio to carry the stereo difference information. Fourier analysis of QAM In the frequency domain, QAM has a similar spectral pattern to DSB-SC modulation. Using the properties of the Fourier transform, we find that: [pic] here S(f), MI(f) and MQ(f) are the Fourier transforms (frequency-domain representations) of s(t), I(t) and Q(t), respectively. Quantized QAM [pic] [pic] Digital 16-QAM with example constellation points. Like many digital modulation schemes, the constellation diagram is a useful representation. In QAM, the constellation points are usually arranged in a square grid with equal vertical and horizontal spacing, although other configurations are possible (e. g. Cross-QAM). Since in digitaltelecommunications the data are usually binary, the number of points in the grid is usually a power of 2 (2, 4, 8 … . Since QAM is usually square, some of these are rare—the most common forms are 16-QAM, 64-QAM and 256-QAM. By moving to a higher-order constellation, it is possible to transmit more bits per symbol. However, if the mean energy of the constellation is to remain the same (by way of making a fair comparison), the points must be closer together and are thus more susceptible to noise and other corruption; this results in a higher bit error rate and so higher-order QAM can deliver more data less reliably than lower-order QAM, for constant mean constellation energy.
If data-rates beyond those offered by 8-PSK are required, it is more usual to move to QAM since it achieves a greater distance between adjacent points in the I-Q plane by distributing the points more evenly. The complicating factor is that the points are no longer all the same amplitude and so the demodulator must now correctly detect both phase and amplitude, rather than just phase. 64-QAM and 256-QAM are often used in digital cable television and cable modem applications.
In the United States, 64-QAM and 256-QAM are the mandated modulation schemes for digital cable (see QAM tuner) as standardised by the SCTE in the standard ANSI/SCTE 07 2000. Note that many marketing people will refer to these as QAM-64 and QAM-256. In the UK, 16-QAM and 64-QAM are currently used for digital terrestrial television (Freeview and Top Up TV) and 256-QAM is planned for Freeview-HD. Communication systems designed to achieve very high levels of spectral efficiency usually employ very dense QAM constellations. One example is the ITU-T G. n standard for networking over existing home wiring (coaxial cable, phone lines and power lines), which employs constellations up to 4096-QAM (12 bits/symbol). Another example is VDSL2 technology for copper twisted pairs, whose constellation size goes up to 32768 points. Ideal structure Transmitter The following picture shows the ideal structure of a QAM transmitter, with a carrier frequency f0 and the frequency response of the transmitter’s filter Ht: [pic] First the flow of bits to be transmitted is split into two equal parts: this process generates two independent signals to be transmitted.
They are encoded separately just like they were in an amplitude-shift keying (ASK) modulator. Then one channel (the one “in phase”) is multiplied by a cosine, while the other channel (in “quadrature”) is multiplied by a sine. This way there is a phase of 90° between them. They are simply added one to the other and sent through the real channel. The sent signal can be expressed in the form: [pic] where vc[n] and vs[n] are the voltages applied in response to the nth symbol to the cosine and sine waves respectively. Receiver The receiver simply performs the inverse process of the transmitter.
Its ideal structure is shown in the picture below with Hr the receive filter’s frequency response : [pic] Multiplying by a cosine (or a sine) and by a low-pass filter it is possible to extract the component in phase (or in quadrature). Then there is only an ASK demodulator and the two flows of data are merged back. In practice, there is an unknown phase delay between the transmitter and receiver that must be compensated by synchronization of the receivers local oscillator, i. e. the sine and cosine functions in the above figure.
In mobile applications, there will often be an offset in the relative frequency as well, due to the possible presence of a Doppler shift proportional to the relative velocity of the transmitter and receiver. Both the phase and frequency variations introduced by the channel must be compensated by properly tuning the sine and cosine components, which requires a phase reference, and is typically accomplished using a Phase-Locked Loop (PLL). In any application, the low-pass filter will be within hr (t): here it was shown just to be clearer. Quantized QAM performance
The following definitions are needed in determining error rates: M = Number of symbols in modulation constellation Eb = Energy-per-bit Es = Energy-per-symbol = kEb with k bits per symbol N0 = Noise power spectral density (W/Hz) Pb = Probability of bit-error Pbc = Probability of bit-error per carrier Ps = Probability of symbol-error Psc = Probability of symbol-error per carrier [pic]. Q(x) is related to the complementary Gaussian error function by: [pic], which is the probability that x will be under the tail of the Gaussian PDF towards positive infinity.
The error rates quoted here are those in additive white Gaussian noise (AWGN). Where coordinates for constellation points are given in this article, note that they represent a non-normalised constellation. That is, if a particular mean average energy were required (e. g. unit average energy), the constellation would need to be linearly scaled. Rectangular QAM [pic] [pic] Constellation diagram for rectangular 16-QAM. Rectangular QAM constellations are, in general, sub-optimal in the sense that they do not maximally space the constellation points for a given energy.
However, they have the considerable advantage that they may be easily transmitted as two pulse amplitude modulation (PAM) signals on quadrature carriers, and can be easily demodulated. The non-square constellations, dealt with below, achieve marginally better bit-error rate (BER) but are harder to modulate and demodulate. The first rectangular QAM constellation usually encountered is 16-QAM, the constellation diagram for which is shown here. A Gray coded bit-assignment is also given.